The digital phase shift keying of a sinusoidal signal (PSK) is one of the most efficient modulation techniques, both in terms of noise immunity and required bandwidth. Nevertheless, the demodulation of PSK signals requires complex demodulator systems. Therefore, other less efficient digital modulation schemes are usually preferred because of their simpler demodulation, for instance Frequency Shift Keying (FSK) or Amplitude Shift Keying (ASK).
The simplest PSK signal is the Binary PSK signal (BPSK). In this case, the carrier phase is shifted between two possible states, 0° and 80°, according to the bit stream. BPSK signals can be easily obtained by multiplying the carrier by +1 (0° phase state) or by −1 (180° phase state). From the receiver point of view, it is impossible to know if the phase of an incoming BPSK signal corresponds to 0° states or to 180° state. This is due to the fact that the actual propagation path from the emitter to the receiver is usually unknown. To avoid this indetermination, the information to be transmitted is coded as transitions between phase states, instead of being coded as fixed phase values. Therefore, when logic “1” has to be transmitted then the phase of the carrier signal is shifted, whereas the phase is unchanged for logic “0”, or vice versa. The signal coded in this way is known as Differential BPSK (DBPSK). It should be noted that from the signal point of view there is no difference between BPSK and DBPSK. The only difference between them is the pre-processing (at the transmitter side) or post-processing (at the receiver side) of the base-band signal. FIG. 1 shows the generation of a BPSK or DBPSK signal as the product of the Base-band signal (derived from the bit stream or from the processed bit stream) and the sinusoidal carrier at the desired frequency.
The usual procedure for demodulating BPSK signals is that of coherent demodulation. Basically, the demodulation process consists of multiplying the received signal by a reference signal at the same frequency as the original carrier.
Mathematically, the BPSK signal can be expressed by:BPSK=±A cos(wt+ψ)  (1)
Where the + sign corresponds to the 0° phase state and the − sign to the 180° phase state. A is the amplitude of the received signal, and ψ is the arbitrary phase due to signal propagation.
The reference signal, S, is given by (the amplitude is set to 1 for simplicity):S=cos(wt)  (2)
The product, P, can be expressed as follow:P=±A cos(wt+ψ)·cos(wt)=±A/2 cos(ψ)±A/2 cos(2 wt+ψ)  (3)
Finally, by low pass filtering P, the following base band term is obtained:PLPF=±A/2 cos(ψ)  (4)
The result is a signal, PLPF, which reproduces the original modulation (±). From (4), if the propagation phase ψ is 0° or 180°, the efficiency of the demodulation process reach its maximum (regardless of the phase indetermination). On the contrary, if ψ=+90°, the efficiency of the demodulation process is null. This fact points out the first drawback of the coherent demodulation of PSK signals, which is the propagation phase uncertainty. The second, and most important, is the availability of a reference signal at exactly the same frequency as the original carrier.
The usual way to overcome both problems is by using a carrier recovery circuit. Carrier recovery is accomplished by using synchronization loops. The most widely used are the squaring loop and the Costas loop, which characteristics and operation are depicted in FIGS. 2 and 3, respectively.
As shown in FIG. 2, the squaring loop consist of a squaring block and a Band Pass Filter (BPF), which from the BPSK input signal generates a reference signal at twice the frequency of the original carrier and, ideally, without any phase modulation. A Phase-Locked Loop (PLL), consisting of a phase/frequency detector, a loop filter and a Voltage Controlled Oscillator (VCO), is used to recover the carrier at twice the frequency. The original carrier is finally recovered using a divide by 2 frequency divider. Demodulation is accomplished by multiplying the recovered carrier by the incoming BPSK signal.
The Costas loop circuit consists of two mixers, which produce the product of the incoming signal with two reference quadrature signals (0°/90°). A third mixer, acting as phase detector, generates an error signal as the product of the low pass filtered outputs of both previous mixers. Finally, the error signal is passed through a loop filter (i.e. an integrator) to generate the control signal of the Voltage Controlled Oscillator (VCO) which, when combined with the 90° phase shifter, generates the reference quadrature signals, and closes the loop. The error signal will be zero when the frequency of the reference quadrature signals is equal to the frequency of the original carrier. Moreover, the VCO output reference signal (in-phase signal) will have either the same propagation phase of the carrier, ψ, or differ from it by 180°. In the locking state, that is to say when the error function is zero, the Costas Loop acts as a demodulator of BPSK signals. In fact, the base band modulator signal (regardless of sign uncertainty) is found at the output of the first low pass filter (LPF1 in FIG. 3).
The main advantage of the coherent demodulation performed by both previous schemes is the tracking of the input signal. This allows the correction of frequency deviations, for instance those due to relative movement between emitter and receiver in a mobile system. Moreover, no previous information about the modulating signal is required (i.e. the bit period). However, synchronization time is usually large, leading to loss of data at the beginning of a communication or malfunctioning in burst mode transmissions. Another important drawback of the synchronization loops is the need of loop filters, which are hard to implement in monolithic form.
In the way of an example, U.S. Pat. No. 5,347,228 employs the coherent demodulation procedure, which is based on the Costas Loop (as shown in FIG. 3), and complemented by a series of additional components for detecting the demodulator tuning status (phase tuning and correct demodulation of the input signal), or the pseudo-tuning status (incorrect modulation).
U.S. Pat. No. 4,631,486 proposes an alternative procedure to achieve a phase reference which permits demodulation. In this case a certain average of the received phasors is carried out, from which a phase reference estimate is obtained. Each received phasor is compared with the reference to demodulate the signal and is then used to refine the phase reference estimate. This procedure possesses the advantage of being able to correctly demodulate signals received in a discontinuous fashion, without loss of information associated with the tuning time. Its inconvenience is the greater complexity of the demodulator system and the implicit requirement to know the modulating signal bit period in order to perform phasor averaging.
Another possible demodulation procedure for signals employing digital phase modulation is the proposal in U.S. Pat. No. 4,989,220. This method is applicable to digital phase modulated signals which only involve changes between adjacent phase states. Basically, the operating principle consists of multiplying the signal received at a time period with the signal received in a previous time period. The time difference is obtained through the use of a delay component and is adjusted so that it is equal to the bit time. The result of this multiplication is filtered by a low pass filter in order to produce the DC component of the resultant signal. Only when there are phase changes in a bit period will there be a change in the value of the DC component. In this case, demodulation is carried out directly, synchronization not being required. The basic disadvantage is that the modulating signal bit period must be known beforehand.